text:
|Publication number
|Publication type
|Application number
|Publication date
|Aug 31, 1993
|Filing date
|Jul 30, 1992
|Priority date
|Jul 30, 1992
|Fee status
|Publication number
|Philip H. Sutterlin
|Original Assignee
|Echelon Corporation
|Export Citation
|BiBTeX, EndNote, RefMan
|Patent Citations (3), Non-Patent Citations (1), Referenced by (76), Classifications (18), Legal Events (5)
|External Links: USPTO, USPTO Assignment, Espacenet
The present invention relates generally to the field of amplifier circuits; more specifically, the invention relates to a class of amplifier circuits for transmitting communication signals over lines primarily used for power distribution.
In advancing the design of communications networks, a trend is developing toward utilizing existing AC power lines as a communications medium. There is a significant advantage to the use of AC power lines for communication purposes. The installation cost for adding dedicated communication wiring and hardware to existing buildings and structures can easily exceed the cost of the system hardware itself. This provides a strong motivation to use existing power lines for communication purposes, as well as for power distribution.
In the past, there has been limited acceptance of power line communication (PLC) systems. Since power lines were never originally intended to provide a medium for communications, there are numerous problems and impairments which must be overcome in order to develop a reliable communication system utilizing existing power lines. Very often, these problems have led to inconsistent and unreliable performance in previous PLC systems.
One of the major problems which must be overcome by a PLC system is the ability to deal with the numerous noise sources and significant signal attenuation which arise in the power line environment. It is well known that most electronic and electrical products (e.g., household appliances) generate significant electrical noise which is eventually coupled back across the power lines. Attenuation of the communications signal results from the frequently lengthy connections and varying impedances seen at the receptacle points. Practitioners in the communications art have long recognized that the combination of signal attenuation plus high noise levels poses an especially difficult communication problem.
As will be seen, the present invention provides a relatively simple, low-cost drive amplifier adapted for power line communication systems. The invented amplifier circuit provides a low output impedance during transmit mode, and a high impedance looking back into the amplifier while operating in a receive mode. Special control circuitry is included to permit rapid switching between transmit and receive modes. Numerous other features desirable of a power line communications amplifier are also incorporated.
Accordingly, it is an object of the present invention to provide an improved power line communications amplifier circuit which provides a low output impedance in the transmission band during signal transmission, and a high input impedance while in a receive mode.
It is another object of the present invention to provide an amplifier circuit for power line communication systems which provides for a relatively large output voltage swing even when driving a very low receptacle impedance.
It is yet another object of the present invention to provide a stable amplifier circuit for power line communications which withstands oscillation even when driving widely varying receptacle impedances.
It is still another object of the present invention to provide an amplifier circuit for power line communications exhibiting low harmonic distortion.
It is a further object of the present invention to provide an amplifier circuit which is compatible with input and control circuitry powered from a integrated circuit operating off a standard power supply.
It is a further object of the present invention to provide an amplifier circuit for power line communications which is capable of switching between transmit and receive modes very quickly to reduce the potential for signal collisions on the line.
It is yet another object of the present invention to provide an amplifier circuit which complies with FCC regulations governing power line communications.
It is still a further object of the present invention to provide an amplifier circuit incorporating all of the above features at a low cost.
In accordance with one embodiment of the present invention, the amplifier circuit includes a driver means for providing a communications signal as a transmit signal. The transmit signal is coupled to a filter means for filtering the signal to eliminate frequency components at a predetermined frequency or frequency range. A buffer means then transmits the filtered transmit signal onto the power line medium through a coupling circuit. The coupling circuit couples the signal onto the powerline while preventing power distribution frequencies from being passed back to the amplifier.
A control means coupled to the buffer means and driver means provides switching from the transmit mode of operation to receive mode of operation. When switched in a receive mode of operation, the buffer means and the driver means present a relatively high impedance to the power line.
One of the central features of the present invention is that all of the elements of the amplifier circuit are combined in such a way as to provide high power driving capability when transmitting, and a relatively high input impedance when in a receive mode of operation, all at a minimal device count and cost.
The present invention will be understood more fully from the detailed description given below and from the accompanying drawings of the preferred embodiment of the invention, which, however, should not be taken to limit the invention to the specific embodiments shown, but are for explanation and understanding only.
FIGS. 1A and 1B show a circuit schematic diagram of one embodiment of the drive amplifier of the present invention.
FIG. 2 is a plot illustrating the output impedance of the amplifier of FIG. 1 (operating in receive mode) as a function of frequency.
FIG. 3 is a simplified circuit schematic of the amplifier of FIG. 1 showing an alternative feedback configuration.
FIG. 4 is a simplified circuit schematic of the amplifier FIG. 1 illustrating the advantage of the feedback configuration.
FIGS. 5A and 5B show a detailed circuit schematic implementation of the amplifier of FIG. 1.
FIG. 6 is an alternative embodiment of the drive amplifier of the present invention.
FIG. 7 illustrates the basic filter topology incorporated within the amplifier of FIG. 6.
An amplifier circuit for driving communication signals onto a power line medium is described. In the following description, numerous specific details are set forth in order to provide a thorough understanding of the invention. It will be obvious, however, to one skilled in the art that the present invention may be practiced without these specific details. In other instances, well known circuits, structures, etc., have not been shown in detail in order avoid unnecessarily obscuring the present invention.
Referring to FIGS. 1A and 1B, there is shown a circuit schematic diagram of the amplifier circuit of the present invention. In transmit mode, the communications signal is applied to node 27. From there, it is coupled through resistor 26 to the positive input terminal of operational amplifier (op amp) 23, i.e., node 28. Phase compensation and gain for op amp 23 are set by resistor 24, coupled between the negative input terminal of operational amplifier 23 (i.e., node 20) and ground, and also by means of the parallel combination of resistor 21 and capacitor 22 coupled between nodes 29 and 20. Additional phase compensation is provided by the network comprising the series combination of resistor 31 and capacitor 33 coupled across nodes 29 and 35 in parallel with resistor 30.
FIGS. 1A and 1B also show diode 36 and resistor 37 being coupled in series between node 35 and the negative supply potential, VEE. Node 35 is also coupled to the collector of NPN bipolar transistor 40. The emitter of transistor 40 is coupled directly to VEE while the base of transistor 40, i.e., node 38, is coupled to VEE through resistor 39. Node 35 is also coupled to the base of NPN bipolar transistor 59. Together, diode 36 and transistor 59 function as a current mirror, with resistor 30 (R1) acting as the input resistor to the mirror. Resistor 37 (R2) and resistor 60 (R3), coupled from the emitter of transistor 59 to VEE, establish the current ratio between the input and output of the mirror. Thus, this ratio sets the voltage gain from node 29 to node 57 in conjunction with the ratio of current mirror load impedance (i.e., the AC equivalent load on the collector of transistor 59) to the current mirror's total input impedance (i.e., resistor 30 in parallel with the series combination of resistor 30 and capacitor 31). This aspect of the present invention will be discussed in further detail shortly.
FIGS. 1A and 1B also show a transmit/receive control signal, TX(ON), being applied to the base of PNP transistor 14 at node 10. The emitter of PNP transistor 14 is coupled to positive supply potential VDD through resistor 12 while the collector of transistor 14 is coupled to the base of transistor 40 at node 38. The base of bipolar transistor 14 is also coupled to the base of transistor 41 through resistor 16. Note that nodes 10 and 17 have a path to ground provided for them through respective resistors 11 and 19.
NPN bipolar transistor 41 is shown having its emitter grounded and its collector coupled to the base of PNP transistor 47 at node 49 through resistor 42. The series combination of resistor 45 and diode 43 connect the positive supply potential VCC to the base of transistor 47 at node 49. The emitter of transistor 47 is coupled to VCC through resistor 46. It should be apparent that the combination of diode 43, transistor 47 and resistors 42, 45 and 46 form a current mirror in the same way described above. As before, the ratio of resistors 45 and 46 establishes the current ratio between the input and output of the mirror.
Notice that the values of resistors 42 (R1), 45 (R2) and 46 (R3) are the same as those of resistors 30, 37 and 60, respectively. This means that the DC current of the upper current mirror, supplying the drive current for NPN transistor 52, matches the DC current of the lower current mirror, which supplies base drive current for PNP transistor 61. Because both current mirrors drive their respective halves of the output stage (comprising transistors 52 and 61) the circuit is balanced with respect to op amp 23 (providing of course that transistor 41 is on, i.e., TX(ON) is high).
Continuing with the description of the circuit, NPN transistor 52 is connected to the collector of transistor 47 at node 53. The base of transistor 52 is also coupled to the base of PNP transistor 61 through the series connection of diode 48, resistor 55 and diode 56. The emitters of transistors 52 and 61 are coupled through biasing resistors 66 and 67.
It is appreciated that transistors 52 and 61 are arranged in a push-pull configuration wherein the collectors of transistors 52 and 61 are coupled to supply potentials VCC and VEE, respectively. Since the current mirrors driving transistors 52 and 61 are balanced, this means that the push-pull output stage is itself balanced. That is, if op amp 23 outputs zero volts at node 29, then the output at node 63 will also be zero. A positive output from op amp 23 causes more drive current to be drawn from transistor 61 relative to transistor 52 so that node 63 becomes more negative. Similarly, if op amp 23 produces a negative output, transistor 52 will receive more base current compared to transistor 61, and node 63 will become more positive.
Feedback is provided for in the circuit of FIGS. 1A and 1B by means of resistors 70 and 71 and also capacitor 72. Resistor 71 is connected between the emitter of transistor 52 (node 64) and the positive input terminal of operational amplifier 23. Likewise, the emitter of PNP transistor 61 is also coupled to node 28 through resistor 70. Resistor 70 is coupled between nodes 65 and node 28. Capacitor 72 is shown being connected across nodes 63 and 28.
Recognize that transistor 59 provides voltage gain for the amplifier circuit at the same time that the push-pull output stage provides current gain. Both of these stages operate to drive a communications signal across a pair of resistors 75 and 76. Resistors 75 and 76 are coupled in parallel across nodes 63 and 80. Node 80 is coupled to one winding of transformer 90 through AC coupling capacitor 83. Transformer 90 provides an interface between the driving amplifier circuit of the present invention and the power lines. Transformer 90 is shown having a 3:1 turns ratio.
Practitioners familiar with PLC systems will realize that in certain situations, signal attenuation can be minimized by the use of various power line coupling networks. When used in conjunction with the present invention, such a coupling network would be connected on the line side of transformer 90. A coupling circuit for use in PLC systems is described in a co-pending application entitled, "Power Line Coupling Network", Ser. No. 07/698,618; filed May 10, 1991, which application is assigned to the assignee of the present invention.
Having thus far described the various connections and devices utilized in the amplifier circuit of FIGS. 1A and 1B, a discussion of various operating principles and advantages of the present invention may now be presented.
One of the important features of the present invention is its ability to transmit a communications signal at the carrier frequency with a low output impedance (e.g., less than 10 ohms). In transmit mode, a very low output impedance is desired in order to overcome the very low impedance of electrical appliances connected to the line. For example, in one embodiment, transmissions take place at carrier frequencies of 118 kilohertz and 134 kilohertz.
Not only must the driving amplifier provide a low impedance in transmit mode, but it also must provide a relatively high output impedance (greater than 500 ohms) at the carrier frequencies while in a receive mode. One can imagine that if a great many amplifiers were connected to a common power line, and if each had a low output impedance in its receive mode, then the collective load impedance as seen by a transmitting source would be so low as to make it prohibitively difficult to transmit a signal onto the line. As discussed further below, the present invention achieves a relatively high output impedance in receive mode and a relatively low the line. As discussed further below, the present invention achieves a relatively high output impedance in receive mode and a relatively low impedance in transmit mode. Moreover, the invention is capable of rapidly switching between the transmit and receive mode with corresponding switching of input impedance levels as seen from the power line. This latter ability is critical for maximizing communications throughout while minimizing signal collisions on the power lines.
During normal transmitting operations, the transmit signal is applied to the positive input terminal of operational amplifier 23 via node 27. The amplified transmit signal appears at node 29. This signal is then coupled to node 35 at the base of NPN bipolar transistor 59. As discussed above, transistor 59 provides voltage gain for the signal as it is delivered to the output stage.
It is significant to note that the portion of the circuit represented in FIG. 1B is powered by the supply potential VCC, which, by way of example, represents a relatively high positive potential of approximately +12 volts (VEE =-12 V). At the same time, the portion of the circuit represented in FIG. 1A is powered by VDD (e.g., 5 V). Thus, the relatively small amplitude signal output by amplifier 23 eventually is gained up to a larger amplitude signal by transistor 59. This larger signal is output onto the line by the push-pull output stage comprising transistors 52 and 61. This stage provides current gain for the output amplifier signal.
Whenever a communication signal is being transmitted onto the power line medium, the transmit control signal, TX(ON), is raised to a high logic level (e.g., 5 volts). A high potential at node 10 turns off transistor 14. At the same time, transistor 40 is also turned off--its base-emitter junction being shunted by resistor 39. Turning off transistor 40 allows the voltage at node 35 to swing freely.
With TX(ON) raised to 5 volts, a high potential also appears at node 17, thereby providing drive current to the base of transistor 41. Because transistor 41 is on, current flows through resistors 45, 42 and diode 43. The resulting voltage drop at node 49 turns on PNP transistor 47 which, in turn, provides a source of base current to activate the push-pull output stage comprising transistors 52 and 61 as explained earlier.
Diodes 48 and 56, together with resistor 55, insure that an adequate voltage separation is maintained between nodes 53 and 57 so that at least one of the two transistors 52 and 61 are operating in the linear or active region at all times. During transmission the output impedance appearing at node 63 is kept to a minimum by means of the negative feedback provided to operational amplifier 23 through resistors 71 and 70. Basically, the output impedance of the amplifier is reduced inversely proportional to the loop gain of the amplifier and the feedback circuit.
The circuit is switched to a receive mode by taking the transmit control signal to a low logic level (e.g., ground). Grounding node 10 activates PNP transistor 14. With PNP transistor 14 on, drive current is supplied to the base of NPN transistor 40. This turns transistor 40 on, thereby pulling node 35 down to negative supply potential VEE, which turns off NPN transistor 59.
Referring now to the upper half of the amplifier circuit, a ground potential on node 10 also grounds node 17. This turns off transistor 41 which causes a high potential at node 49 turning off transistor 47. With transistors 47 and 59 both off, the push-pull output stage comprising transistors 52 and 61 are disabled, i.e., the transistors have no drive current. Disabling the push-pull output stage results in a high output impedance at node 64 and 65. Hence, grounding the control signal TX(ON) causes the amplifying portion of the circuit of FIGS. 1A and 1B to switch to a high impedance state. In this state the received communication signal is simply tapped off of the amplifier side of transformer 90 (node 94).
As previously discussed, the invented amplifier is capable of switching between transmit and receive modes of operation very quickly--with typical speeds on the order of 15 microseconds. The ability to rapidly switch between one mode of operation and the other is crucial to the avoidance of collisions between signals on the power line medium. As the switching delay gets larger, the probability of incurring collisions increases dramatically. For example, according the CSMA protocal, a node sensing carrier holds its transmission until it detects a quiet line; however, if the drive amplifier takes an inordinate amount of time to settle, then it takes that much longer for the other nodes to become aware of another transmission.
To achieve fast switching, several important circuit features have been incorporated in the present invention. First of all, providing an input node which can be driven digitally by standard logic devices is important in providing a control element for switching operations. All of the components shown in FIG. 1A can be easily integrated into a single semiconductor circuit using conventional bipolar complimentary metal-oxide-semiconductor (BiCMOS) process technology. Alternatively, a conventional CMOS process can be used with appropriate substitution of transistors 14, 40 and diode 36. This means that the switching circuitry of the amplifier is easily interfaced to the switching control signal TX(ON). Moreover, when the circuit portion illustrated in FIG. 1A is fabricated as an integrated circuit, all the control and input circuitry can be powered off of a common 5 volt power supply.
It should be understood that when the circuit portion of FIG. 1A is fabricated as an integrated circuit, that there will be a corresponding shift in supply voltages. For example, one possibility is VDD =5 volts, VEE =OV and VCC =+24V. Furthermore, it is appreciated that some resistor values may need to be adjusted (e.g., resistors 42, 45 and 46) to accommodate this shift in supply voltage.
Secondly, the values of resistors 11, 12 and 39 are chosen to insure that transistor 14 always operates in a mode where it is either functions as a current source or it is completely turned off. Transistor 14 is deliberately kept out of saturation so that node 38 can be switched quickly from a high to a low voltage level. Furthermore, the presence of resistor 11 allows rapid discharge of the base voltage of transistor 14 during a high to low transition of node 10. Correspondingly, the presence of resistor 39 helps in the rapid discharge of the voltage at node 38 to turn off transistor 40. Resistor 19 performs the same function to discharge node 17, coupled to the base of transistor 41, when the control signal TX(ON) transitions from a high to a low level. Altogether, the DC biasing keeping transistor 14 out of saturation and the inclusion of resistors 11, 19 and 39 facilitate rapid switching between modes.
One of the difficulties in designing an amplifier circuit for driving power line communications is the fact that government regulations limit the amount of power that may be broadcast at certain frequencies. According to the present invention, the power delivered to the line is kept relatively high without harmonic levels which would exceed government regulation limits. This is accomplished by means of transistor 59 and push-pull transistors 52 and 61, and the configuration wherein the relatively small output of op amp 23 is stepped up to a larger signal. As explained previously, transistor 59 provides voltage gain while current gain is provided by the push-pull combination. At present, the output stage of the invented amplifier circuit is capable of delivering about 40 milliamps with a communications signal of 18 volts peak-to-peak at node 63.
Recognize that the operating supply potential of the output driver stage illustrated in FIG. 1B is currently maintained at approximately 24 volts (e.g., �12 V). Because of the relatively large currents and voltages which must be withstood by the components of FIG. 1B, each of the component elements illustrated in FIG. 1B is implemented as a discrete device. The combination of higher voltage discrete transistors (represented by the components of FIG. 1B) driven by an integrated circuit providing mode switching and signal amplification, minimizes the total cost of the entire circuit. The operational amplifier and switching portions of the circuit may be implemented on an integrated circuit as will be discussed further. Meanwhile, the high-powered output portion is optimally manufactured at a discrete level. Taking this manufacturing approach, the total cost of the circuit of FIGS. 1A and 1B is kept to a minimum.
Another advantage of the present invention is its ability to operate off of very small current (tens of milliamps) from the power supply, even under worse case conditions. Drawing such a small amount of current from the power supply allows the present invention to utilize a capacitive input power supply. A capacitive input power supply is well known to power supply designers as a low cost supply. The prohibitive factor associated with the capacitive input type of power supply is that it is limited in the amount of current it can deliver before the capacitor size and cost make them impractical (generally not exceeding 50 milliamps). However, it compensates for this fact by its ability to generate higher voltages which translate into high power output (recall that power is equal to the product of voltage times current). By minimizing the current drawn from its supplies, the present invention is able to benefit from the use of a capacitive input type of power supply. That is, the present invention relies upon a relatively large operating supply potential (e.g., 24 volts) compensated by a relatively low supply current (approximately 40 millilamps), to achieve maximum power output.
This result is accomplished by means of the large voltage gain provided by transistor 59 coupled with the large current gain of the output stage, and by the 3:1 turns ration associated with transformer 90. It is appreciated that transformers not only transform impedance but they also modify voltage levels. By way of example, for transformer 90 of FIG. 1B, the voltage level on the amplifier side is nominally three times larger than the voltage level present on the line side. But at the same time, the transformer provides three times the current drive capability on the line side, with the benefit of smaller supply currents on the amplifier side. The present invention employs large operating supply potentials at the output stage in combination with the 3:1 turns ration of transformer 90 to transform the high voltage associated with the amplifier side of the coil into a low voltage on the line side. (Obviously, other turns ratios may be used with similar results.)
On the other hand, current draw from the amplifier's power supply is minimized by means of the 3:1 turns ratio. That is, the current drive capability on the line side is three times greater than the current drawn from the power supply on the amplifier side of transformer 90. The low power supply currents then allow the use of the inexpensive capacitive input type of power supply.
Because an amplifier driving a power line communications system experiences widely varying receptacle impedances, assurances must be taken to prevent oscillations from occurring. This is often a difficult challenge in amplifier design. To provide stability against oscillations, the present invention incorporates lead/lag compensation networks coupled to op amp 23. The lead compensation network circuitry comprises resistor 31 and capacitor 33 coupled in series between nodes 35 and 29. Lag compensation is provided by the feedback network comprising resistors 21, 24 and capacitor 22. At the frequency where the gain of the amplifier plus the gain or loss of the feedback network is unity, the lead and lag compensation networks guarantee at least 45� of phase margin.
FIG. 2 illustrates the input impedance looking in from the power line of the circuit as a function of frequency. By way of example, operating frequencies of 118 kilohertz and 134 kilohertz are shown being utilized for a particular embodiment. The impedance is depicted by waveform 91. The lower impedance at frequencies below 118 kilohertz is primarily due to the magnetizing inductance of transformer 90. At frequencies above 134 kilohertz, the low impedance is due to the presence of parasitic capacitance.
One of the reasons why the amplifier circuit of the present invention is useful in driving power line communication systems is that it achieves a very low harmonic distortion. Basically, FCC Class B government regulations prohibit conducting more than 1000 microvolts of noise across any portion of the AM broadcast band. Emissions tests are performed to meet these regulations by use of a network which results in approximately 100 ohms across the hot and neutral power lines in the pertinent frequency range. The power lines are then measured to determine how much noise is being passed onto the lines. Because of the operating frequencies involved (e.g., 118 KHz and 134 KHz), there is a potential for a fifth harmonic to fall within the AM broadcast band.
Low harmonic distortion is attained in the present invention by enclosing the entire amplifier circuit--not only just the operational amplifier itself, but the operational amplifier as well as the output buffer circuitry--within the feedback loop. In other words, the feedback connection back to the positive input of operational amplifier 23 is taken from the output node of the amplifier driver circuit at node 63. This feedback path from node 63 to node 28 includes resistors 66, 67, 70, 71 and capacitor 72. The fact that the feedback connection is around the whole amplifier--including the output buffers, instead of simply operational amplifier 23 alone--means that the present invention realizes distortion levels of better than 68 dB below the fundamental when driving a 100 ohm resistive load.
Furthermore, amplitude distortion (i.e., signal clipping) is similarly avoided when driving low receptacle impedances on the order of 1 ohm. Consider the simplified alternative embodiment shown by the circuit diagram of FIG. 3. Rather than employing separate resistors 70 and 71 coupled to the emitters of transistors 61 and 52, respectively, the schematic of FIG. 3 shows the feedback resistors merged into a single equivalent resistance 92 connected to node 63. With this type of feedback, the open loop output impedance of the amplifier is approximately RB /2. (Note that the purpose of resistors 66 and 67 is to provide a stable DC quiescent current for the two output transistors 52 and 61). With the loop closed, the output impedance looking into node 63 is approximately 0 ohms. The equivalent resistance between node 63 and 80 is shown in FIG. 3 by resistor RO.
One possible problem with the alternative feedback scheme of FIG. 3 is that when the output stage has to drive a very small load impedance (one ohm or less) a significant amount of extra current is generated. This extra current drops additional voltage across resistors 66 and 67 which limits the voltage clipping level at node 63. In other words, the peak-to-peak voltage swing of the signal appearing at output node 63 must be reduced to avoid incurring amplitude distortion. The consequence of this extra drop means that less power is delivered onto the communication lines.
Contrast the alternative embodiment of FIG. 3 with the embodiment shown in FIG. 4, once again in simplified form. Instead of making the feedback connection directly to node 63, the embodiment shown in FIG. 4 has feedback connections to nodes 64 and 65; that is; directly to the emitters of the output devices. Also, the single feedback resistor of FIG. 3 is shown being split into a pair of feedback resistors 70 and 71. By splitting the feedback resistor into two separate resistors and coupling them directly to the emitters, the amplifier is able to reduce the value of the resistance of RO by RB /2 for the same output impedance seen looking in from the power line.
This result is significant since it allows the resistance value between nodes 63 and 80 to be reduced in value. So in effect, a portion of the previous resistance RO has now been embedded into the amplifier. The net result is that an additional 2 dB of output signal range is achieved before clipping occurs. In other words, the key difference between the schematics of FIGS. 3 and 4 is that in FIG. 4, the series output resistor between node 63 and 80 can be made smaller by RB /2 to provide extra signal headroom. It is appreciated that the resistance RO, i.e., resistors 75 and 76 as shown in FIG. 1B, is required to keep the amplifier operating in a linear mode when presented with a very low impedance on the line. Linear operation guards against voltage or current clipping.
FIGS. 5A and 5B are a detailed circuit schematic diagram showing one implementation of the present invention including specific component types and values. Note that the circuit of FIGS. 5A and 5B includes several EXCLUSIVE-OR logic gates together with extra associated circuitry. Practitioners in the art will appreciate that the purpose and function of this circuitry is to eliminate possible glitches which might occur while switching from transmit to receive mode and vice-a-versa. Of course, certain embodiments and applications may not require such circuitry, or may employ other components for similar reasons. This additional circuitry is not deemed to be essential to the present invention.
An alternative solution to the problem of driving communications signals onto power lines is provided by the circuit schematic of FIG. 6. Like the previous embodiments, the amplifier of FIG. 6 provides a means for presenting a high impedance to the power line while operating in a receive mode. On the other hand, when transmitting, the amplifier is characterized by a very low output impedance. In addition, digital control circuitry is provided to facilitate rapid switching between transmit and receive modes of operation.
To begin with, the amplifier of FIG. 6 has an output stage which comprises a unity gain buffer 175 coupled between a supply voltage VA (e.g., 9 volts) and ground. Buffer 175 provides an output signal at node 159 which is then coupled through capacitor 180 to a suitable power line coupling circuit. For ease of understanding, the amplifiers' output signal VO is shown across a load resistance RL which represents the equivalent impedance of the coupling circuit and the power line.
Unity gain buffer 175 includes a pair of up/down transistors configured to provide large amounts of current into output node 159. At the same time, this configuration produces a relatively high impedance looking back into the amplifier from the power line when switch 115 is in the "off" position. The up/down configuration in the top half of buffer 175 comprises the combination of PNP transistor 137 and NPN transistor 168. Both of these transistors are configured as emitter followers, with the emitter of transistor 137 (through resistor 190) and the base of transistor 168 both being coupled to node 139. The emitter of transistor 168 is coupled through series resistors 160 and 161 to output node 159. These resistors have relatively small resistance values so that the signal voltage drop across them is negligible. This means that the potential at node 133 (coupled to the base of transistor 137) is essentially the same as the potential produced at node 159 during normal transmit operations.
Similarly, NPN transistor 136 has its base coupled to node 133 and its emitter coupled to node 138 (through resistor 193). Node 138 is coupled to the base of PNP transistor 169, which has its emitter coupled to output node 159 through series resistors 162 and 163. Transistors 142 and 122--together with resistors 141 and 124--provide a source of current for driving buffer 175.
Altogether, components 115-124 and 141-146 comprise a biasing network coupled to unity gain buffer 175. When digital switch 115 (which may comprise an ordinary field-effect digital gate) is in its "on" position, supply potential VDD is coupled to node 120 through resistor 116. This establishes a reference voltage on the bases of transistors 121 and 122. Collectively, devices 116-124 act as a current mirror supplying bias current to transistor 136 and the other current mirror comprising devices 141-146, which in turn provide bias current to transistor 137. Devices 141, 142 and 145, 146 function as a current mirror such that a constant current is provided for transistors 136 and 137. Thus, with switch 115 in its "on" position, unity gain buffer 175 is activated.
In the opposite case, when switch 115 is in its "off" position, node 120 is grounded. This turns off transistors 121 and 122 and also transistor 142. The result is that no drive current is available for buffer 175 and the emitter follower configuration of the up/down transistor pairs simply presents a high impedance to the power line. Practitioners in the art will appreciate that with switch 115 "off", the unity gain buffer configuration shown in FIG. 6 presents a very high impedance looking back into node 159 (ignoring diodes 151 and 152 for the time being). This up/down configuration also provides high current driving capability so that the invented circuit is capable of transmitting a signal of approximately several hundred milliwatts of power.
Diodes 151 and 152 are incorporated into the basic buffer circuit of FIG. 6 in order to prevent high currents from destructively breaking down the output stage transistors. During its receive mode of operation, the amplifier of FIG. 6 must be capable of withstanding unattenuated signal voltage swings which appear on the power line. These voltage swings could possibly destroy the output stage transistors by means of high reverse base/emitter breakdown current. To guard against this possibility, diodes 151 and 152 are included.
During normal transmit operation the voltage on node 133 is approximately equal to the voltage on node 159, so that no current flows through either diodes 152 or 151. In receive mode of operation, however, buffer 175 is disabled and the voltage VO can swing across the full operating potential. But as soon as the voltage at node 159 deviates from the voltage at node 133 by more than ˜0.7 volts, one of the two diodes 151 or 152 will begin to conduct. Thus, potentially destructive reverse breakdown current is shunted away from the output bipolar transistors.
High impedance at node 133 (as seen looking back toward the input) is further preserved by means of a digital-to-analog converter (DAC) comprising a standard binary-weighted string of resistors coupled in parallel to convert a digital representation of the transmit signal into an analog signal at node 111. The DAC resistors are driven by a DAC driver circuit 100 which provides an appropriate voltage to the respective DAC lines, e.g., DAC0-7. Driver 100 also includes circuitry for synchronizing the individual input lines of an 8-bit input digital signal to a clock signal. In a current embodiment, DAC driver 100 comprises part no. XC3042PC84-100 manufacured by XILINX Corporation.
The use of a digitally-controlled input signal is important in the present invention because it provides an additional means of establishing a high receive mode input impedance at node 133. DAC 100 has a thevenin equivalent impedance that represents the parallel combination of all of its resistors. Because the circuit of FIG. 6 is driven digitally, this digital circuitry can be tri-stated to create a high impedance looking back into node 133. Thus, a high output impedance is achieved in the "off" or receive mode of operation by debiasing the emitter follower output stage by means of digital switch 115, and also by tri-stating the digital drive circuitry of DAC 100. At the same time, diodes 151 and 152 perform the dual function of protecting the emitter follower base emitter junctions from reverse breakdown in receive mode while limiting the output current to protect against shorts while in transmit mode. By way of example, if RL =0 (i.e., a short), then buffer 175 would attempt to supply an infinite current at its output --limited only by the available base drive current for transistors 168 and 169 the beta of these transistors and the very low impedance of resistors 160-163. On the other hand, the inclusion of diodes 151 and 152 provides a more properly controlled current limit, wherein the source and sink peak currents through transistors 168 and 169 are limited to approximately 0.7 V/R160 and 0.7 V/R163, respectively.
In order to prevent transients from appearing at the amplifier's output while switching from transimit to receive modes (or vice-a-versa) the cumulative change on capacitors 102 and 104 is conserved. For example, consider the case where the circuit is switched from a transmit to a receive mode of operation. When operating in transmit mode (and just prior to switching) node 101 and the line labelled DAC7 are both charged to one-half of the supply potential VDD (e.g., 2.5 V). In receive mode, however, (after switching) resistor 103 pulls the left plate of capacitor 104 up to VDD, while the line labelled DAC0 is pulled low. This grounds the left plate of capacitor 102. Thus, the cumulative charge on both capacitors is conserved. (Note that line DAC0 is the only DAC output line which is not tri-stated in receive mode of operation.)
Note that the quiescent current IDQ is selected in the circuit of FIG. 6 such that there is sufficient base drive current available for transistors 168 and 169. In general, resistors 190 and 193 are selected such that the sum of resistors 160-163 ratioed against the sum of resistors 190 and 193 set an appropriate output quiescent current IOQ which achieves low distortion and meets FCC regulations.
Finally, the power line amplifier circuit of FIG. 6 also incorporates a important filter function for maintaining low distortion in the transmit signal. Low distortion and spurious filtering is critical if the power line amplifier is to meet stringent FCC regulations. Accordingly, the embodiment of FIG. 6 includes a Salen Key filter having a circuit configuration which is best illustrated by the schematic of FIG. 7. The Salen Key topology utilizes a unity gain buffer in conjunction with capacitors C1 and C2 and resistors R1 and R2, to provide a wide variety of filter functions. In the present invention, a second order filter is realized having a cut-off frequency point at approximately 400 KHz. This operation is in accordance with the current use of the circuit of FIG. 6 in a spread spectrum communication system operating within the frequency range of approximately 100-400 KHz. A third order Salen key filter can be realized by including the additional elements RX and CX as optionally shown in FIG. 7.
Viewing FIGS. 6 and 7 together, it can be seen that capacitor C1 comprises the sum of capacitors 127 and 126, capacitor C2 comprises capacitor 131, and resistor R2 comprises resistor 130. Resistor R1 of FIG. 7 comprises the parallel combination of resistors 109 and the seven other DAC resistors (i.e., R, 2R, 4R, 8R, 16R, 32R and 64R; collectively, RDAC) in FIG. 6. Resistors 107 and 108 set the amplifier's bias boltage at one-half the supply voltage VCC so that a balanced plus/minus swing is available.
One of the noteworthy features about the circuit of FIG. 6 is that it combines a second-order filter function with a "tri-stateable" unity gain output buffer wherein both share common components. Obviously, combining these two functions in a single circuit reduces the overall component count. The end result is an amplifier circuit providing a complex filter function while at the same time having the capability to drive large amounts of power onto the power lines at a minimum component cost.
Whereas many alternations and modifications to the present invention will no doubt become apparent to a person of ordinary skill in the art after having read the foregoing description, it is to be understood that the particular embodiments shown and described by way of illustration are in no way intended to be considered limiting. Therefore, reference to the details of these embodiments is not intended to limit the scope of the claims which themselves recite only those features regarded as essential to the invention.
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|Oct 7, 2014
|Conversant Intellectual Property Managment Incorporated
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|The Boeing Company
|Digital communication over 28VDC power line
|Sep 11, 2002
|Mar 31, 2005
|Patrick Sullivan
|Coupling circuit arrangement for data communication over power lines
|Aug 3, 2006
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|Amplifier circuit of BTL system
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|Mar 19, 2010
|Sep 30, 2010
|Mosaid Technologies Incorporated
|Telephone outlet for implementing a local area network over telephone lines and a local area network using such outlets
|Jul 29, 2010
|Nov 18, 2010
|Cooper Technologies Company
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|Jul 29, 2011
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|Ac-powered in-wall computing device with power-line networking capabilities
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|Aug 31, 2007
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|Method and apparatus for a power line communications test system
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|Maxim Integrated Products
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|Elcom Technologies Corporation
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|U.S. Classification
|330/51, 330/296, 340/310.17, 330/306, 340/310.15, 340/310.13, 307/3, 330/267, 330/297, 340/12.38, 340/12.34, 340/12.36
|International Classification
|Cooperative Classification
|European Classification
|Jul 30, 1992
Owner name: ECHELON SYSTEMS, CALIFORNIA
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST.;ASSIGNOR:SUTTERLIN, PHILIP H.;REEL/FRAME:006219/0324
Effective date: 19920723
|Feb 27, 1997
|Fee payment
Year of fee payment: 4
|Dec 30, 1997
|Certificate of correction
|Feb 27, 2001
|Fee payment
Year of fee payment: 8
|Feb 28, 2005
|Fee payment
Year of fee payment: 12


text:
Designer DependenciesWhen you are designing screens in Ignition you are constantly linking (binding) properties to tags and other properties on the same window. Each time you bind a property to something else you are setting up a dependency.
Let's look at a small example. Add a tank component and a slider component to a window. Bind the tank's value property to the slider's value property through a "Property" binding. By doing that, the tank is now dependent on the slider. If you try to delete the slider component Ignition won't let you saying "Cannot Delete. Non-Selected component Root Container.Cylindrical Tank depends on selected component Root Container.Slider."
People always ask, "Is there a way I can see who is linked to who?" Yes, the designer has a built-in feature to show all of the dependencies. Just simply click on View > Dependencies > Show All from the file menu in the designer. It will draw arrows from components to other components to illustrate such dependencies.
The direction of the arrow is important. In our example above, the arrow is drawn from the slider to the tank. The arrow shows how data flows. When the slider changes the tank will update. So, if the arrow is going away from the component it means those components are linked to it.
This is an extremely important feature when designing templates or indirect popup windows. Components inside of templates are always linked in some way to the template parameters. That way the template can dynamically change. With indirect popup windows you pass in a value that is used in all of the component property bindings.
In both cases, if you turn on the dependencies you can easily see if one of the components is not linked to the template parameter or the value passed in. If you don't see line drawn to the center something may be wrong.
|Pressure label doesn't have an arrow because it is not bound
using the template parameter. In this case that is expected.
Designer SpotlightsSpotlights are another great feature of the designer. Spotlights allow you to see invisible components, components that have bindings, and components that have scripting on the window by painting a box around the component.
There are a lot of situations where you will have invisible components in the designer. It can be challenging to figure out if there are any and where they are. If you turn on the invisible spotlight, it is easy to pinpoint where they are. Click on View > Spotlights > Invisible to turn the spotlight on. The image below shows that there are two components invisible (the purple outline) in this window.
If you turn on both features dependencies and spotlights, you will get a good idea of how a window is configured.

subset: commoncrawl


text:
This event occurs weekly, on Sunday.
Town Center Corte Madera celebrates summer with the annual Summer Music Series, set to take place every Sunday, June through August. Free to attend, everyone is invited to get rockin' to live musical performances in the beautiful outdoor setting of Town Center. June kicks-off the series with an exciting lineup including a mix of pop, rock, and blues from The Eddy Shin Group; soulful tunes from The Growiser Band; guitar beats from The Fourth Son; and West Coast folk rock and Americana from Solid Air.
We make every effort to ensure the accuracy of this information. However, you should always call ahead to confirm dates, times, location, and other information.


text:
- Place of birth:
nottingham, nottingham, United Kingdom
- Date of passing: Feb 27, 2011
- Place of passing:
nottingham, chilwell, United Kingdom
|Let the memory of Chloe be with us forever
A beautiful young talented girl, taken from us so soon. Her voice was the sound of an angel, easy enough to make you drift of into the stars. A loving daughter to barbara and her brother cleston. Our thoughts go out to her family, and us friends. she was the sort of person to make you laugh no matter how sad or happy you are, chloe was always the one to cheer us all up. we've all shared many memories with her, from primary school and secondry, il always remember her as the little girl who used to run round sunnyside school play ground with your ringlets bobbin up and down :( fly high in the sky baby, save atleast 1000 places for all of us. awww what was you like ayyy, always got somert daft to say but ud still make us burst out into laughter. what a heart of gold she had, R.i.p Baby, You'll Never Be Forgotten Always In Our Hearts Forever & Always, we love youuu <3<3<3
"6 years today chloe, i am sorry no one has wrote in a while but always know we are thinking about you, i know i am i listen to your music and remember in my ears the soft beautiful sound of your voice i know your in a better place than this horrible cruel world but i still cant help but wish you was here with us, maybe im being selfish or maybe im right in wishing but all i know is if you are finally happy then i must be too i look into the clouds and smile hopeing you are smiling back. You made everyone smile and happy even though you wernt i wish we seen underneath yours before it was too late. I miss you girl so i lay this flower just for you because even after 6 years my wish wont come true.
Rest in piece beautiful Angel.
Let your voice sing with the gods."
Our beautiful talented girl!
We all miss you so much I hope you're partying and showing everyone up there what you're made of!
It's still not any easier knowing God gained an angle so early on in her life!
I'm sure you're so proud of everyone and how there making the lives!
Rest in paradise sweet pea!
Love you always"
on the 27th of February it was 3 years since you passed away and im so sorry im only wrighting this now but i just got my laptop back..i miss you so much and these past 3 years hasnt been the same without you i hope your alright up there and that you are singing your little heart out..well you must be because there are lots of bright sunny days and i know when we have sunny days its because your watching down on us and your singing, i miss you so much your gorgeous smile your amazing voice your kind and loving nature just everything and even 3 years gone past its still so weird you not being here i think of you everyday and its haunting me that your not here but i will see you soon babe hope your alright and i love and miss you so much and so does everybody else .. you will never ever be forgotten so rest in piece princess and fly high and keep singing!!! <3 <3 <3
"Didn't know you but I know you was a lovely girl and you was an amazing singer you had a golden heart and a lot of friends fly high beautifiul xxxxx"
sat here listening to you sing! i miss you so bad :(.. i hope you're up there doing what you do best making people laugh singing youre soul out and being the girl you always have been!! you were amazing chlo' and every one misses you like crazy
R.I.P beautiful always love and always missed xxxxxxxxxx"
"missing you chloe
fly high babe
never be forgotten
r.i.p chloe hayden <3"
"r.i.p chloe you beautiful girl i cant believe you are gone i miss you so much kind, funny, beautiful, bubbly girl with a heart of gold i dont know if there is a heaven or not but all i know is if there is then god obviously was jealous by how much you made everyone around you happy and how amazing you were that he wanted you for himself and didnt want to share you..never forgotten babe <3"
"You were amazing! Beautiful, talented and had such a warm heart.
You made all of our days bright with your bubbly attitude.
Theres so much stuff that i wish i could say but im never going to get the chance. I hope your doing well up there, You'll always be in my heart. <3"
"My Unforgettable Niece R.I.P."
"youll always be missed, rest in peace <3"
"ever since chloe passed away we have had bright sunny days just proves she had such an amazing voice keep singing to those angels chloe you have done everyone so proud :) xx"
"Foughts go out to close friends and family to chloe as it was her funeral today and i no it would of been tearful day for everyone! Wish i could of been there to pay my respects and to say goodbye but thanks to my dad he didnt txted bk :( anyway R.I.P CHLOE HAYDEN you will be missed loads by everyone! Sleep tight angel <3. <3."
"Oh and someone has to put one of her songs on this website ! Let the world hear her soul"
"Chloe, I didnt know you personally or very well at all, but I do remember seeing you around school in Chilwell, you were the year above me. You used to have singing lessons with Ronnie, and Id hear your voice. Your voice. You have a beautiful voice and God will keep you safe. I am so sorry to hear of your loss, I pray for you and your family. I hope the angels are with you. Love Chloe McShane xxx"
"R.I.P chloe you will never be forgotten, i hope you are singing with the angels because your voice is amazing.you was so pretty and a bubbly girl i can't believe your gone too soon.sleep tightxx"
"will be sadly missed. I remember Chloe as such a lovely, kind, fun person who always wore a smile on her face.
You will always be remembered. we know you'll never be forgotten XXX" chloe"
"R.I.P Chloe you will always be loved and never forgotten. You was so talented and it could of took you far in life. You was always so bubbly and cheerful and it was only last week when i saw you in beeston :( <3"
"you where one of my mates threw primary school and i saw you and talked to you on the street when i saw you i really cant believe yove gone R.I.P angel <3 xxxxxxx"
"R.i.p Chloe You beautiful girl,
going to miss seeing you around with your big smile , i didn't believe it when i first heard and i still don't, it seems too unreal to be you, I'm going to miss you. Oh and you should of seen how many of your friends was at the bandstand for you, i got you some flowers :) your missed already, my lovely childhood friend xxxxxxxxxxx"
"So sad to hear about Chloe. Such a wonderful young woman who will be sadly missed. I remember Chloe as such a lovely, kind, fun person who always wore a smile on her face.
You will always be remembered.... sleep tight Chloe, we know you'll never be forgotten XXX"
"A beautiful talented girl, and from someone who hasnt seen you for a long time my heart is aching so bad. Always a happy bubbly little girl at sunnyside running around causing trouble with my sister and paige :') you sleep tight sweetheart, il share a prayer for you. Never forgotten lots of love Tevra & Rhea xxxxxxxxxxxxxxxxxxxx"
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text:
- Age / Gender:
- n/a, Female
- Location:
- Under the sky
- Joined:
- 8/17/09
- All Stats >
I want to be an artist, but I dabble in music and such. I'm your average nerd. =) Any animation I do is generally a .gif or .mov since I don't actually have flash.
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text:
DVD of the Week – Samurai Rebellion (1967), Considered in Connection with Harakiri (1962)
by HELEN GEIB
Director Masaki Kobayashi's Samurai Rebellion is a heartrending drama about a family destroyed by its clan. It is a companion piece to Kobayashi's Harakiri (1962) and with that film, a devastating indictment of feudal Japan and samurai culture. Unlike Harakiri with its complex flashback structure, Rebellion is a traditional linear narrative. It does not allow its audience the luxury of emotional respite as it moves inexorably toward its tragic conclusion.
The family in Rebellion is the patriarch Sasahari Isaburo (Toshiro Mifune), his son Yogoro, Yogoro's wife Ichi, and the young couple's baby daughter. While the family structure is almost a mirror image of the family in Harakiri, the difference in their situation is significant. The family in the earlier film lost their place in the social hierarchy upon the dissolution of their clan. They fall victim to a callous society and the casual cruelty of a clan to outsiders. In contrast, the family in Rebellion is betrayed by its own clan when they refuse to accede to the leader's morally unconscionable and legally indefensible demand that Ichi be given to him as his mistress.
Both films explode the myth of samurai honor, Harakiri with a story told from the perspective of the clan outsider and Rebellion from that of the insider.
Both films also provide pointed commentary on the status of women in samurai society. The father in Harakiri turned down an offer to give his daughter to a wealthy man as a concubine. Selling his daughter would have secured lifelong financial security and a respectable social position for them both, and his refusal is greeted with uncomprehending disgust by the intermediary.
Rebellion pushes the issue squarely to the forefront. Isaburo and Yogoro are the only members of the Sasahari extended family who object to the clan leader's demand. The rest of the family, including Isaburo's wife and younger son, are eager to comply in equal hope of currying favor and avoiding reprisal. Ichi's wishes are consulted only after the men have proved obdurate in their refusals, and questions to her of what she wants are invariably accompanied by threats of the deadly consequences of refusal.
Samurai Rebellion is available on DVD from The Criterion Collection. Features include a new digital transfer and newly translated subtitles, an interview with Kobayashi, an essay by Japanese film historian Donald Richie, and the theatrical trailer.
New releases this week: Married Life, Outsourced, The Promotion, Then She Found Me
Sep 4 2008
Another film that touches on some of these themes is Samurai Gold Seekers AKA Sword of the Beast AKA Kedamono no Ken.


text:
Today, we begin our week long celebration of our THANKSGIVING HOLIDAY. It is truly a joyful time of the year to be blessed and bless others as God has purposed us to do. So come on in and share with us each day as we share with you God's word for His Blessings over your life and the lives of all His Children of the World. The World is in turmoil and it needs all of us to do our part in helping our brothers and sisters to be lifted up and stand tall in the world that we share as one in Jesus.
Together we will celebrate this thanksgiving in a special way, seeking God's blessing for all he has done to allow us this opportunity to share with each other our abundance from the feast of the harvest. Remember, we are our brothers keeper, therefore we must reach out to those that are in need of food and shelter, offer them a helping hand and then watch how God will add to our blessings.
The Bible is filled with ways for you to give thanks and praise God from cover to cover. The choice is yours to take the time to study and understand, the many reason why you should always be giving thanks. The Psalms provide us with a rich history of how we can do it. Take some time and study these Psalms and use them this thanksgiving with your family and friends as you share a thanksgiving meal with them.
Do it with your mouth, not just your mind – Psalm 51:15, 63:5, 66:8,66:17, 71:8, 15, 24, 105:2, 109:30, 119:171-172, 145:21
Seek God, thirst for Him – Psalm 22:26, 42:2, 63:8, 105:4
Thank God continuously – Psalm 71:6, 8, 14-15, 75:9, 79:13, 92:1-2,115:18, 145:2
Tell people about God's wondrous works – Psalm 26:7, 96:3
Thank and praise God in front of your brothers and sisters in the congregation, among many people- Psalm 22:22, 25, 26:12, 35:18,68:26, 111:1, 116:19
Among the unsaved of all nations, thank and praise God – Psalm 18:49, 57:9, 96:10, 108:3
Praise God to the next and all generations – 79:13, 102:18, 145:4
Sing praises to God, sing with understanding, and make it glorious –Psalm 7:17, 47:7, 66:2, 66:4, 9:2, 27:6, 28:7, 57:9, 61:8, 63:7, 68:4,32, 69:30, 71:22-23, 92:1, 96:2, 101:1, 104:33, 105:2, 135:3, 146:2
Even sing for joy in your bed Psalm 149:5
Do it with all your heart – Psalm 9:1, 86:12, 111:1, 138:1
Be glad and rejoice in God – Psalm 9:2
Fear God, honor Him, glorify Him, revere Him – Psalm 22:23
Love and respect God's house – Psalm 26:8, 138:2
Don't sin, seek righteousness – Psalm 33:1
Play musical instruments to praise God, and do it skillfully – Psalm 33:2-3, 43:4, 71:22, 92:3, 149:3, 150:3-5
Shout for joy, rejoice – Psalm 33:3
Be creative in your praise (write or sing God a NEW song, not always the same old one) – Psalm 33:3, 149:1
Stand in awe of Him – Psalm 33:8
Remember and praise God's names – Psalm 44:8, 45:17, 54:6
Lift up your hands to Him – Psalm 63:4
Think about God at night – Psalm 63:6
Encourage other to praise and thank God – Psalm 67:3-5
Praise him with dancing – Psalm 150:3, 30:11, 149:3
After reading the Psalmists' instruction, it would be wise to ask ourselves: Do our lives truly express thankfulness to God? Or are we numbered among those are living a life of ingratitude, failing to act like we understand who God really is and what he has done?
Related articles
- A Psalm of Thanksgiving. (greatriversofhope.wordpress.com)
- Psalms 67. A Song Of Thanksgiving (biblestudytruthtoday.wordpress.com)
- Psalm 138. A Prayer Of Thanksgiving. The LORD's Goodness To The Faithful (bummyla.wordpress.com)
- Offering Spiritual Sacrifices. (greatriversofhope.wordpress.com)
- Continually thanking and praising God (teresasthoughtsfortoday.wordpress.com)
- Psalms 67. A Song Of Thanksgiving (bummyla.wordpress.com)
- Blessing and Thanking God through the Pain and Devastations of Life (chronicillnesspaindevotionals.wordpress.com)
- Five Thanksgiving Day Commands. (greatriversofhope.wordpress.com)
- What are You Thankful to God For? (drwendycampbell.wordpress.com)


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Johnstone/StarPix/INFphoto.com, Anthony, PacificCoastNews.com
Julia Louis-Dreyfus Skips 2018 SAG Awards, Wins for Outstanding Performance by a Female Actor in a Comedy Series for Veep
Johnstone/StarPix/INFphoto.com, Anthony, PacificCoastNews.com
Warning! Round 7 of our Ho Ho Ho Awards may make you feel a burning sensation...down there.
These two regulars on the dating scene always find somebody new to hook up with, 'tho we'll never understand why. Both slew-bags treat pretty much everyone around them like crap:
Date Perplexed Ho: Tila Tequila vs. John Mayer
Tila Tequila Li'l Miss MySpace babe came out as a lesbian—there goes every fella's shot at love with her. But we were moreso confused when she disgustingly took Rihanna down a notch by blogging that she's Team Chris Brown (among other nasty stuff). This is after she supposedly got roughed up by ex-BF football player Shawne Merriman; she even leaked the pics of her bruises to the press herself. Way to be a ho to your own kind, Tila. Let's hope she tames the ef down if she really is pregnant.
John Mayer: Stayed with Jennifer Aniston long enough to get to go to the Oscars—then dumped her, twice. Everything this schmuck does to women—bitching to the press and writing obvious songs and blog posts about his exes, especially—just reeks of classlessness. For somebody so obsessed with ladies, he sure never has learned how to treat them.
Did John Mayer have the year's best song? Check our Best of 2009: Killer Tunes and find out!


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Homemade face packs are commonly prepared by many women. If you want to get glowing and flawless skin naturally, you can apply face packs made with simple ingredients. These ingredients can be fruits, vegetables and spices. Yes, fruit face packs are commonly seen everywhere. Even fruits like lemon, strawberries, banana and papaya are used to make face wash. However, getting a face pack made with vegetables is a strange beauty regime. There are many women who apply face packs made with vegetables like carrots, potatoes, avocado etc.
These face packs are very effective too. Want to try some homemade face packs made with vegetables? Check out...
Carrot face pack: The underground vegetable is rich in vitamin A. It is good for sensitive skin and has been used as a natural remedy to cure acne, pimples, whiteheads, pigmentation and sun tan. Blend carrot into juice and add few drops of honey, lemon juice and milk. Apply on your face and massage for 2 minutes. Leave for 15 minutes. Rinse with cold water. You can also massage on dark patches of skin around knees or elbows to whiten it. You can also mix carrot juice with orange
Tomato face pack: Mash a tomato into puree. Add few drops of lemon juice, ½ tsp multani mitti (Fuller's Earth) and sandalwood powder. Mix well and apply this vegetable face pack on your face and neck. Let it dry. Wash with cold water to get glowing skin. There are many other tomato face packs that you can try at home.
Potato face pack: This is another face pack that is made with a vegetable. If you have dry skin, you can mash a potato and massage on your face to moisturise it and prevent it from itching. Another way of applying potato is my grating it and then mixing with curd. Mix well and apply on the face. Leave for 10 minutes and then rinse with milk and cold water.
Avocado face pack: This vegetable hydrates the skin, moisturises it and prevents it from drying. During winters, you do not need to rely on oily winter creams. Just blend a small piece of avocado with orange peel. Add 1tsp yoghurt and egg white. Mix well to form a paste. Apply it on your face and neck. Leave for 10-15 minutes. Rinse with cold water to restore radiance to dry and sensitive skin.
Onion face pack: Did you know that the onions that make you cry have many skin benefits? You can apply onion face pack to get glowing, flawless and acne-free skin. Onions have vitamin A, C and E. These vitamins protect the skin from sun damage and also reduces free radicals from the body. In a bowl, add ½ tsp Bengal gram flour, sandalwood powder, curd, lemon juice and onion paste. Mix well and add few drops of milk to make a paste like face pack. Apply on the face and leave for 10 minutes. Rinse with cold water. This homemade face pack exfoliates the skin, removes dead skin cells and also lightens the skin.
These are a few homemade face packs made with vegetables. Do you know more vegetables that can be applied on face?


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Christina Aguilera Long Straight Formal Hairstyle - Light Blonde (Platinum) Hair Color
Description(Added: 27th June 2012 )
This light blonde hairdo is blow-waved smooth with body added to the roots and movement to the ends to achieve this fantastic 'do perfect for any special occasion. This look is easy to re-create at home with the right tools and products and great to frame a round face.
- Hair Category: Formal
- Hair Length: Long
- Hair Elasticity: Straight
- Gender: Women
- Face Shape: Round, Oval, Heart, and Triangular
- Hair Density: Medium/Thick
- Hair Texture: Fine/Medium
- Age: Under 21, 21 - 30, 31 - 40, and 41 - 50
- Height: Any
- Weight: Thin/Average/Large
- Glasses: Suits with and without
- Styling Time: 30+ minutes
- Products: Mousse, Wax, Smoothing Shine, and Hair Spray
- Styling Tip: Use your finishing products to smooth down the ends and direct your hair into place.


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This introductory 10 video course will teach you how to evaluate your energy in four different ways. You learn them all but then choose your favourite for each circumstance.
Once you have learnt to master the four techniques of evaluating your energy, you'll be shown the 3 tiered system of evaluating the life energy. This quick and simple system will give you the freedom in finding out what your body is trying to tell you so you can finally give it what it needs.
This introductory workshop will give you the basic tools and fundamental skills to further your study in the field of Energy Medicine.


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Series: Elm Creek Quilts Novels
Part: 3
(From the publisher):
Although they share a common creative objective, the Cross-Country Quilters find their friendship tested by the demands of everyday life. Yet despite differences in age, race, and background, the women's love of quilting and affection for one another unite them. The quilt they create becomes a symbol of the threads that hold their lives together-a glorious patchwork of caring and loyalty that brings home an enduring truth: Friends may be separated by great distance, but the strength of their bond can transcend any obstacle.(from the publisher)
Original title: The Cross-Country Quilters
Original languages:
Genre: Fiction→ General Fiction


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For the longest time I was not a fan of any type of lip gloss until one day I changed my mind and decided to try the L'Oréal Paris Glam Shine Stain Splash and I liked it. I'm also a sucker for anything YSL and have had my eye on the glossy stains for a while and I had quite a lot of Boots points to use, so I treated myself to it. Both lip products have been known to be quite similar, I'm going to share with you my opinions on them, if they really are alike and worth the money.
L'Oréal Paris Glam Shine Stain Splash
YSL Rouge Pur Couture Vernis À Lèvres Glossy Stain
A little bit pricey at £22.50 and it is expected from a designer lip product. As per usual YSL have beautiful packaging, I think girls are a lot like magpies, we like pretty, shiny things. I went for a deeper, darker colour in number 10, it is a red with a hint of magenta. I'm just going to say it, I love it. The colour is just what I was looking for, there are also a wide range of colours available. Most importantly it lives up to its name of 'glossy stain' it is a nice non-sticky gloss. I have to be careful when applying it because of how well it stains the lips so I don't accidentally smudge onto my skin, I would say it lasts me over 4 hours, I have not actually counted but one day I applied it at around lunch time, topped up once and the colour was still there when I came home in the late evening. I have found no problem with it making my lips feel dry, overall it is great.
I hope this has been useful to anyone who is thinking of purchasing either product.
xo Natasha


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Under some circumstances, the fault tolerance (FT) driver cannot initialize after a failure of a mirrored boot partition (containing Windows NT system files) and system partition (containing NTLDR and boot loader files). This article provides a step-by-step system recovery procedure for such a failure.
More Information
There are two recovery options:
Option 1
- If the hard disk drive with the primary system partition has failed and you have a single controller mirror set established, set the physical SCSI ID on the mirror drive to zero. If you have a duplex mirror set, swap the drive from the primary controller to the secondary controller.
- Use a Windows NT Fault Tolerance (FT) boot floppy disk to point to and boot the system/boot partition. Make sure the BOOT.INI file points to the partition with the Windows NT system files.
- Open Disk Administrator, break the mirror set and mark the primary system partition on Disk0 as Active so that the Windows NT FT disk is not necessary for the next startup.
- If the failed drive has been replaced, establish the mirror set and allow data regeneration during the next system boot.
Option 2: No Hardware Changes
- Before breaking the mirror set in disk administrator, shut down the server and use the Windows NT Fault Tolerance boot floppy disk to point to and boot the remaining healthy partition (the mirror).
- Open Disk Administrator, select the mirror set partition and break the mirror. The healthy partition retains the drive letter previously assigned to the mirror set. The faulty partition, if it is still available, is assigned the next available letter.
- Delete the faulty partition on Disk0 or replace the disk drive if necessary. You cannot delete the active boot partition through Disk Administrator in 80x86-based computers; you have to use the Windows NT Setup Disk to delete it.
- Establish the mirror between the healthy system/boot partition and the raw disk space on Disk0. Exit Disk Administrator and save the disk configuration changes.
- Use the Windows NT FT boot floppy to start regenerating the mirror set to Disk0. Time required for this depends on factors such as disk size, access time and controller type.
- Verify that the mirror set is healthy and has completed regenerating, then select the set and break the mirror. You have to do this on 80x86- based computers because the system partition needs to be marked as Active for startup, and Disk Administrator allows you to mark only primary partitions on Disk0 as Active.
- From Disk Administrator, modify the partition drive letters to have the appropriate assignments (C drive on Disk0 and D drive on Disk1). Mark the primary partition on Disk0 as Active.
- Reboot the system without the Windows NT FT boot floppy. Run Disk Administrator and delete the partition on Disk1 and then re-establish the mirror from Disk0 to Disk1.
رقم الموضوع: 120227 - آخر مراجعة: 27/03/2007 - المراجعة: 1


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You will need the following items for this spell:
- no ingredients
- no thing spell
- no needs spell
spell atracts males or females by love
Casting Instructions for 'love spell'
please listen to my tries
I invoke thee
I call on you to make girls love me
this is my will
so mote it be
|Print Spell
|Mark this spell as Spam
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text: Pregnancy in humans is generally considered to be 9 calendar months or 40 weeks from the time of conception. These 9 months are divided into 3 periods called trimesters that are each roughly 3 months long. Each trimester has specific points of growth and development for the fetus. A defined full-term pregnancy is generally at 40 weeks but after 36 weeks a pregnancy is usually considered to be full-term.


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Joelle Aeschlimann from ECAL created an ingenious iPad accessory called 'Little Boxes'. This delightful little device has a crank on top and interacts with iPad to create something magical. It's hard to describe in words what it exactly does so the best way to do so is to just watch the demo video to get a feel for what it does.
So in essence, Little Boxes are music boxes that you can hand crank to create certain visual and sound effects. It was created using Adobe Air for iPad . It will not work on your iPhone simply because the device is too big for the iPhone.
Tutor: Alain Bellet, Gael Hugo, Christophe Guignard
ECAL / University of Art and Design, Lausanne Switzerland
Bachelor Media & Interaction Design


text: Awards Entertainment Conference Recognizes Steve Levitan with Commitment to California Award Advocacy State of Entertainment Industry Highlighted at Hollywood Chamber Conference Chamber Hollywood Brings Back Holiday Lights to Tinseltown! Welcome to Hollywood Ripley's Believe It Or Not! Odditorium Share: Museums Ripley's Believe It Or Not! Odditorium Visit Website 6780 Hollywood Blvd Hollywood, CA 90028 (323) 466-6335 Business Directory Upcoming Events New Members Click here to make a payment: Chamber Videos View all videos Premier Sponsors


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CORD, Jubilee MPs in Custody Over Hate Speech Claims And Law Breaking
Eight administrators yesterday spent the night in police cells as the State moved into take action against contempt discourse.
The administrators needed to persevere through coercive capture, a three-hour confinement in dim cells and a further detainment of four days in remand after police requested more opportunity for examinations.
They are presently are gazing at a Sh1 million fine or a correctional facility term of up to three years if sentenced. Their captures and arraignment in court was the summit of emotional scenes as police looked for them save for Celebration coalition's Moses Kuria (Gatundu South) and Ferdinand Waititu (Kabete).
Mr Kuria and Mr Waititu survived first light strikes at their city homes after they showed up at the Directorate of Criminal Examinations to compose explanations on Monday night. They exited DCI workplaces at 9pm in the wake of being without given bond.
Kimani Ngunjiri (Bahati) was additionally not captured after he exhibited himself to the police.


text: Things are looking up as DTS is finishing out the semester. We have had a very productive start to 2010 adding two excellent members to our brotherhood, and making some nice house improvements. Alex Pojman and Kevin Beene are our two newly activated brothers, and scored higher on their tests than anyone in the last few semesters. Both are passionate about DTS and we are looking forward to big things from both of them.