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Metasurface-Based Dual Polarized MIMO Antenna for 5G Smartphones Using CMA

This paper exhibits a low profile dual-polarized MIMO antenna with high isolation to meet the requirements of 5G smartphones. The integration between a vertically polarized slot and a horizontally polarized slot is investigated and applied for 28 GHz dual-polarized smartphone antenna. The antenna is combined with metasurface (MTS) to achieve high gain and more directivity. In order to design the metasurface, the characteristic mode analysis is used to investigate the performance of MTS at 28 GHz. The proposed antenna achieves high isolation coefficients better than 40 dB and cross polarization lower than -40 dB from simulated and measured results. The isolation between elements is achieved without any additional decoupling techniques. The proposed MTS slot antenna operates with 4 GHz bandwidth (26-30 GHz) with a realized gain of 11 dBi and efficiency of 90%. Four antennas (with eight ports) are positioned orthogonally at the corners of the mobile PCB to serve MIMO for 5G applications. The effect of MIMO antenna on the human is taken into consideration in power density term. Furthermore, the housing and components of smartphones are taken into our consideration in this paper.


I. INTRODUCTION
Nowadays, the communication systems have been developed rapidly and attract the researchers due to its new features such as high data rate, wide bandwidth, high resolution, heterogeneous services, virtual networks, provide hugely broadcasting data, and fast action. One of these communication systems is the new cellular generation that is called the fifth generation (5G) [1], [2]. The 5 G system has been planned to provide a gold key to the quickly growing demand for mobile data traffic (data rate), overcoming the constraints of the present capability of communication technologies. Therefore, the International Telecommunication Union (ITU) has established several groups to achieve all 5G standards before 2020 and the ITU release the applicable frequencies for the new mobile generation (5G) between 24 GHz and 86 GHz [3]. Even though the range of 5G still under review, The associate editor coordinating the review of this manuscript and approving it for publication was Giorgio Montisci .
there are several candidate bands [4]. There are different bands such as sub 3 GHz, sub 6 GHz and millimeter bands that are recommended for 5G applications. But many of researchers selected the millimeter bands because the low frequency bands are crowded and overloaded by different applications, therefore the low frequencies can't meet the required broadband of the 5G [1], [5]- [15]. The range from 28 GHz to 38 GHz is highly recommended [16]. The antenna is considered as a mastering key for this new communication system. In order to design an effective antenna for 5G mobile phone, there are several fundamental challenges that need to be considered; one of these challenges is the free space loss (FSL) and atmospheric absorption (AA) that have large values due to the higher frequency of millimeter ranges.
To solve the problems of high losses in MM-wave, substantial efforts have been introduced to design 5G antennas with high gain, small size and novel geometry such as patch [11], slot [14], [17], phased arrays [18], switchable antennas [6], and dipole antenna arrays [19] are presented for miniature the antenna size. All the aforementioned designs are presented for miniaturizing the antenna size but they have their own constraints such as feeding network loss, low gain, and the complexity of the structures.
Nevertheless, most of the antennas in this range are limited to the linearly polarized antennas, while in the real case the mobile terminal will encounter different sorts of movements in Euler areas in addition to the characteristic loss in the millimeter bands. Therefore, the miss polarization among the transmitter and the receiver antennas is one of the main significant loss factors in this communication system. For full utilization of power in 5G systems, the antenna of dualpolarization is a good candidate for solving the problems of power losses and increase the bit data rate of the communication systems [20]. So, the antenna with different polarization (polarization diversity) plays an essential key to solve the aforementioned problems.
In [21] Yang Li et al., introduce a hybrid eight ports orthogonal dual-polarized antenna for 5G smartphones, this antenna consists of 4 L-shaped monopole slot elements and 4-C-shaped coupled fed elements. The 4 L-shaped are printed at the corners and the 4 C-shaped are printed at the middle of a thick 1mm FR-4 substrate. This design achieves 12.5 dB, 15 dB for the isolation and the cross-polarization, respectively. Through the past months, Zaho et al. [22] present a dual-polarized antenna for 5G/WLAN based on the integration between inverted cone monopole antenna and cross bow-tie antenna for VP and HP, respectively. The cross bow-tie antenna is fed by a 90 • phase difference feeding network. Therefore, the separated power divider and phase shifter are used as a feeding network.
In [23] Huang et al. introduce a dual-polarized antenna that consists of a main radiator, an annulus, and a reflector. The main radiator consists of two pairs of differentially-driven feedlines to transmit the energy to the coplanar patch. This structure achieves 26 dB and 35dB for the isolation and the cross-polarization, respectively. Eight-ports dual-polarized antenna array is reported in [24], the proposed antenna array is composed of four square loops and each loop is excited by two orthogonal fed coupled feeding strips.
The multiple-input-multiple-output (MIMO) system is preferred for 5G smartphone applications to meet the high demand for maximizing the throughput and the quality of service. In other words, the MIMO antenna technology is one of the most significant components of future wireless communication schemes as it improves throughput without raising the input power and the bandwidth. However, the incorporation of the MIMO antenna scheme into the same board for handheld devices that have a small size is challenging owing to the high mutual coupling between the adjacent antenna components, particularly when they are spaced less than a half-wavelength apart [25]- [28].
To address the aforementioned problems and challenges/limitations, this article recommends a compact dual-polarized MIMO antenna for handheld 5G devices. The dual-polarization is introduced due to its mastering performance in introducing a solution to improve the spectral efficiency, to enhance the isolation and to enlarge channel capacity. In addition to use of metasurfaces to improve the gain of the proposed antenna. The proposed design consists of two slot antenna: the first is a slot antenna with bottom microstrip feed line and the second is a co-planar slot antenna with CPW feed on the top layer. The 5G design operates at 28 GHz with dual-polarization and achieves high isolation between ports 43dB and cross-polarization (40 dB). The proposed structure has a peak realized gain (5-6 dBi), and radiation efficiency (85-95%). The proposed antenna based on the characteristics mode analysis /theory of characteristics mode is introduced. The MIMO antenna consists of 8 ports to serve the high data rates and the channel capacity of the 5G.

II. ANTENNA DESIGN AND ITS PERFORMANCE A. ANTENNA CONFIGURATION
The proposed antenna consists of two orthogonal slots to achieve pure dual-polarization and to give two diversity patterns as shown in Figure 1. The antenna is implemented on a Roger 4003C substrate with a dielectric constant of 3.38, tangential loss of 0.0027 and a thickness of 0.2 mm. The thin substrate is selected to reduce the losses at millimeter band and to be compatible with the end launch connector (1.85 mm) that has a very thin pin.
The slot antenna is one of the popular antennas that is used for smartphones in last few years [1], [29]- [34] due to its simple structure and its multiple operating modes. Figure 1 (a and b) shows the proposed slot Ant. I. The slot is etched on the ground plane that is printed on the top layer and fed by the 50 ohm microstrip line. The feed line is printed on the opposite side of the ground plane to feed the slot antenna. The second antenna (Ant. II) is a slot antenna fed by CPW line as depicts in Figure 1 (c and d). As a result of the thin layer substrate, the width of the CPW-feeding line is wider than the pin of the end-launch connector (The width of CPW feed line without bottom ground is W f2 = 2.5 mm while the diameter of end-lunch pin is 0.18 mm). Therefore, we used a CPW with a ground plane to decrease the width of CPW-feeding line and to achieve 50-ohm input impedance. After this, the CPW-feeding line is tapered to match the impedance of the slot. The third antenna (Ant. III: proposed antenna) consists of the integration between Ant. I and Ant. II configurations to achieve the dual-polarization from the two antennas. The two feed lines printed on different faces of the substrate as shown in Figure 1 (e and f), furthermore, the feed lines are orthogonal together. Moreover, the two slots are used to prevent the coupling between the two feed lines.  Figure 2 shows the current and electric field distributions of the slot antennas I and II. In the case of Ant. I, the magnitude of surface current on the ground plane are the strongest above the microstrip line, thus the slot executes the highest disruptive impact of this current. The current is completely impeded near the center of the slot and induces a charge build-up on the long sides of the slot that acts as a capacitor. Moreover, the current near to the parties of the slot bends around its ends to continue to pass and this like the inductor. Therefore,  the slot antenna equivalents to two shunt transmission lines shunted by a parallel integration of an inductor and a capacitor. Also, a radiation resistor can be added to the equivalent circuit of slot antenna for long slots.
The proposed slot antenna operates with a length equivalent to 0.5λ g from resonant frequency at the center: where L s is slot length, c is the velocity of free space, f r is the resonant frequency, and eff is the relative effective permittivity of the proposed antenna. Figure 3 shows the reflection coefficient of Ant. I at different values of length and width of the slot. For Ant. II, the wide bandwidth is achieved due to the excitation of multiple resonate modes by the combination of the CPW and the slot. The resonant frequency and BW are tuned by the length and width of the slot (L s2 ,W s2 ). Figure 4 shows the effect of length and width of the slot on the operating bandwidth.  The polarization of Ant. I is vertical and that of Ant. II is horizontal. W s1 and W s2 are the dimensions of slots width to control the matching of the vertical and the horizontal modes, respectively. Furthermore, y 0 is a tuning parameter for matching port 1, and W t , L t are parameters to optimize the matching at port 2. To consider the practical case, we take the end launch connector into account in our designs as shown in Figure 5. Therefore, length of the feed lines are increased by 5 mm in x and y directions to avoid the interconnection between the two connectors. A high isolation between the two ports can be expected due to the orthogonality characteristics of the two ports and symmetric/ antisymmetric characteristics of the three modes of CPW (with the ground). The reflection coefficients and the isolation coefficients of the proposed antenna are shown in Figure 6. One can notice that the isolation coefficients between the two ports have high values through the operating bandwidth (more than 45 dB) and the antenna has a good matching (S 11 and S 22 ). The antenna achieves 2.2 GHz as a wide bandwidth from 27 GHz to 29.2 GHz when port 1 is excited and 5 GHz from 25.6 GHz to 31.6 GHz for port 2. The proposed antenna achieved common bandwidth (2.2 GHz) to cover the 5G applications at 28 GHz.
The Ant. III is redesigned without ground in the bottom (the width of CPW line without ground is recalculated) to make the antenna with only one common ground in the top. The S-parameters of this design is shown in Fig. 7. The results are still ensure that the antenna has good matching and high isolation between its ports. Figure 8 and Figure 9 show the surface current and electric field distributions for the two ports. The surface currents and electric field of the proposed antenna at 28 GHz for the two ports are presented to ensure that the proposed antenna achieves dual-polarization between their ports. It is clear to note that the surface current and the electric field flow along the y-axis when port 1 is fed. While they flow   along the x-axis when port 2 is excited. Therefore, dualpolarization is achieved. The radiation patterns of the proposed antenna from port 1 and port 2 in both XZ and YZ planes at 28 GHz are shown in Figure 10. We can observe that the cross-polarization levels in both planes are less than 40 dB as compared with the co-polarizations. The antenna achieves a gain of 6.23 dBi and 6.85 dBi as shown in Figure 11 when excited from port 1 and port 2, respectively.

III. METASURFACE DESIGN AND ANALYZE A. CHARACTERISTIC MODE ANALYSIS
As described in [35], the characteristics currents are obtained by solving a particular eigenvalue equation that is derived VOLUME 8, 2020 From (2), the characteristic modes (CMs) are introduced based on the coefficient matrix's generalized values by Harrington [36]: where ν n : eigenvalues, R is a real part of the impedance matrix/operator for the MoM, J n is an eigencurrents (Eigen function) and by noting (2) and (3), the characteristic values can be written in terms of eigenvalues as follows, where λ n corresponding characteristic values to the eigenvalues. The eigenvalues have a range from −∞ < λ n < +∞.
To solve the MoM equation, the CMs are used as a basis function to expand the unknown total current, J, on the surface of the metal as where V i n is the excitation coefficient on the conductor surface. The excitation coefficient can be expressed as where E i is the impressed E-field. The first step to the analysis of the CMs is to analyze the eigenvalues because they introduce the data on how the related modes (J n ) radiate and how they are related to the resonance. Therefore, the eigenvalues are used as an indicator to know the resonant frequency for each characteristic mode.
The second parameter for CMA is the characteristic angle (α n ): The characteristic angle calculates the difference in phase between the characteristic currents (J n ) and its related characteristic fields. The characteristic angles values are within the range 0 0 ≤ α n ≤ 360 0 . The third parameter for CMT is a model significance (MS): This term represents the inherent normalized amplitude for each current mode J n and it is named the modal significance. If its value close to 1, the mode meets the resonance condition. From MS equation (8), we are able to calculate the resonance of the CM in addition to the operating bandwidth of CM. The CM that has a resonance must be at λ n = 0 and MS n = 1 and the CM that doesn't contribute to the radiated field is at MS n = 0. The multilayer solver and integral equation solver from CST Microwave Studio are chosen to implement the CMT with very high quality [37]- [39]. These solvers are used to calculate the characteristic modes and those related parameters.

B. DESIGN OF METASURFACE (MTS)
In this section, the analysis of MTS and the integration between MTS and antenna are introduced. The MTS of 7 × 7 unit cells (square patches) is proposed at 28 GHz as shown in Figure 12. The square patches are printed on Rogers RO 4003C substrate with a dielectric constant of 3.38 and a thickness of 0.2 mm. The CMA solver from CST Microwave Studio is used with free space boundary conditions to analyze this structure. To examine the metasurface modal behaviors, the first ten modes are calculated (Eigenvalues, characteristic angles, and modal significances) in the range from 18 GHz to 38 GHz as shown in Figure 13 (a)-(c). Over the band, we notice that only the modes from 1 to 9 that have resonant frequencies are appeared. In this design, the vertical and horizontal polarizations are desired. Therefore, we need two similar modes (one is VH and the other is HP) over the proposed band. In this design, we have two groups of degenerate modes (J 1 /J 2 and J 7 /J 8 ) that can be used, but the other modes are not considered according to (6). On the other hand, J 7 /J 8 are at the end of operating band (31 GHz). Figure 13 (a)-(c) shows that the mode J 1 /J 2 are the only two modes that have pure resonant at 28 GHz. Also, J 1 \J 2 have characteristic angle equal to 180 0 at 28GHz in addition to zero Eigenvalue at 28GHz.
The modal electrical surface currents are shown in Figure 14 (a) and the current directions of each mode are indicated by black arrows. All the fields and currents in this section are calculated at 28 GHz. As can be noticed, first modal current (J 1 ) is in phase through the MTS and its polarization in the y-direction. Also, the second modal current (J 2 ) is typical as (J 1 ) but with 90 0 out of phase. In other words, J 2 is directed in the x-direction through the MTS. Therefore, J 1 /J 2 are a pair of orthogonal modes. As a result of that all currents of the first and the second modes are in phase, they have pure broadside radiation as shown in Figure 14 (b). J 3 and J 5 have symmetrical distribution about y-axis, and x-axis, respectively with null current at the center of MTS and a null along z-axis in the radiation pattern as shown in Figure 14 (b). The current of the third mode (J 3 ) flows as a closed loop with null at the center and thus it looks like the behavior of inductor, which can be verified from its characteristic angle at about 28 GHz is being below 180 0 . The currents of mode J 4 and mode J 6 are self-symmetrical about y and x axis, respectively. J 7 and J 8 are 90 0 out of phase and symmetrical around 45 0 from x-axis and y-axis, respectively. The last two modes have quasi-quadrature symmetric about x-axis and y-axis at the same time. Clearly, the only modes J 1 and J 2 have good main lobe whereas the others modes have split main lobes. Therefore, these are unacceptable modes according to (6) and the theory of mode expansion.

IV. ANTENNA DESIGN WITH MTS
In this section, the MTS is used to provide high gain, wide bandwidth, and compact size and to reduce the back radiation VOLUME 8, 2020 from the slot antenna. The integration between the proposed antenna and MTS is as following: • The dual polarized antenna is based on two slots are introduced to operate at 28 GHz (in section II).
• Then we optimize the dimensions of metasurface unit cell to have the pair of orthogonal modes (J 1 -J 2 ) with broadside radiation. Moreover, the others modes of MTS are out of the focused band and they have spilt main lobe (in section III A).
• In this section, the integration between the dual slots antenna and the MTS is introduced. The slots that are adjusted to operate at 28 GHz are used to excite the modes J 1 and J 2 of the MTS that have pure resonant at 28 GHz. The MTS is feed by the two slot antennas to excite the first and second modes of MTS at 28 GHz. Therefore, the two small slots are etched from the MTS and aligned to the slots of the antenna to increase the coupling between the antenna and the MTS. Figure 15 Figure 16. Figure 17 shows the measured and simulated reflection coefficients and isolation coefficients of the ports for the proposed antenna. The measured operating frequency of the   proposed antenna is 28 GHz which in good agreement with the simulated results. The results confirmed that the proposed antenna achieves wide bandwidth from two ports that satisfy the requirements of the 5G communications in terms of bandwidth. The proposed antenna achieves good isolation between its ports (more than 40 dB). The normalized radiation patterns of the MTS antenna are shown in Figure 18 for port 1 and port 2 in the x-z plane and the y-z plane at 28GHz. The co-and cross-components are introduced with more than  40 dB between them. Furthermore, the MTS achieved low back radiation at the two ports in the x-z plane and the y-z plane.

V. MIMO ANTENNA DESIGN
In our proposed MIMO antenna, the antenna elements are positioned at the corners of the handset board with a total dimensions of 100 × 60 mm 2 as shown in Figure 19 for the design configurations and prototypes of the MIMO antenna with MTS. The simulated and measured reflection coefficients of the 8 ports MIMO antenna are introduced in Figure 20 (a and c) in order to ensure that the MIMO antenna has more than 2GHz bandwidth shared between its  ports. The isolation coefficients related to port 1 are introduced in Figure 20 (b and d). One can notice that the worst isolation coefficient is higher than 36 dB between port 1 and VOLUME 8, 2020   port 2 in the measured case and more than 40 dB in the simulated case. The simulated and measured reflection coefficients of proposed antenna achieve wide bandwidth from two ports (26-30 GHz for port 1 and 25-30 GHz for port 2) with shared bandwidth more than 3 GHz between all ports.
The radiation efficiency of the two ports are illustrated in Figure 21 (a). The efficiency of the proposed MTS antenna is around 92% within the whole band. The gain of the proposed antenna is shown in Figure 21 (b), one can notice that the MTS is used to increase the gain of the proposed antenna and to achieve a gain of 11 dBi. The results from other ports are similar to port 1 and port 2.
Some of the antennas in smartphones require a common ground plane between its MIMO elements. Therefore, the MIMO antenna with printed common ground plane on the top layer is presented in Figure 22. The common ground plane does not have any significant changes on the reflection coefficients of the MIMO elements, in contrast, it reduces the isolation between ports by small significant amount as shown in Figure 23. One can notice that the worst isolation coefficient is higher than 37 dB between port 1 and port 2. Furthermore, all ports have good matching and achieve the required BW for 5G applications.
The envelope correlation coefficient (ECC) is one of the main parameters to evaluate the MIMO performance. The ECC is used to calculate the similarity between the antenna performances and to evaluate the diversity between the elements of MIMO. The acceptable value of ECC should be less than 0.5 [25]- [28]. Whereas the lower values of ECC mean that each of the two antennas are good isolated. The ECC can be calculated based on the radiation pattern as: where ρ: ECC, F(θ, ∅): radiation patterns of antenna #m or #n, m and n are number of the antenna m,n = 1,2,. . ..,8. Figure 24 shows the different ECC values between the MIMO elements (two elements each time) based on the 3-D radiation pattern of each element. The values of ECC is very small due to the different polarization between the neighbour antennas. It is observed that the values of ECC is less than 0.02 and this means that the MIMO antenna has a good diversity performance.

VI. SMARTPHONE MODELING A. IMPACTS OF HOUSING AND PHONE COMPONENTS
To simulate the real environment of the smartphones, the antenna is integrated with the housing and the components of mobile as shown in Figure 25. The screen, speaker, camera, battery, and other components are considered with the proposed MIMO antenna and all components are covered     Figure 26. One can notice that the reflection coefficients from all elements are affected due to the existence of the housing. The reflection coefficient of the ports are slightly shifted but are still have good matching and achieve the requirements for millimeter 5G. On the other hand, there is a high isolation between ports. The 3-D radiation patterns of MIMO elements are presented in Figure 27.We can notice that the radiation patterns are in different directions due to the diversity between the elements.

B. IMPACTS OF HUMAN HEAD
The research on health risk from the electromagnetic waves produced from wireless terminals is introduced in the VOLUME 8, 2020   literature. The Specific Absorption Rate (SAR) is a figure of merit for evaluating the power absorbed by the human tissues for the frequencies used by current mobile communications networks of second, third and fourth generation (2 G, 3 G and 4 G) [40]- [43]. The SAR quantifies are the absorbed energy per unit tissue volume and the SAR values should follow one of two standards: American standard (1.6 w/kg) for each 1 g or European standard (2 w/kg) for each 10g [44]- [47].
For the millimeter wave range, there are two approaches to calculate the electromagnetic exposure to the human: 1) SAR: Some papers in the literature evaluate the electromagnetic exposure by the same previous definition of SAR [5]. 2) Power density: the term to calculate the electromagnetic exposure into the human body changed from SAR to the term of power density (Pd) because the absorption becomes more superficial due to the fact that penetration is very low at higher frequencies [48]- [51]. Figure 28 shows the distribution of SAR from 8 elements for 10g standard. The SAR values for the two standards are summarized in TABLE 4. The antenna is proximity close to the human head model with 0.5 mm distance and inclined as in the take mode by (60 0 ). The reference power of the proposed antenna elements at 28 GHz is set to 24 dBm for each element.   In the second approach, IEEE, FCC and International Commission on Non-Ionizing Radiation Protection (ICNIRP) introduced frequency limits at which the definition of SAR calculation shifts to power density calculation as shown in TABLE 5. The conversion frequency at which this shift in exposure metric is 3 GHz, 6 GHz and 10 GHz, for IEEE, FCC, and ICNIRP, respectively. In other words, at mm-Wave frequencies, PD is currently preferred due to the difficulty of determining a reasonable volume for SAR assessment when the penetration depths are very low [44]- [47]. The power density exposure into the human model is calculated as shown in TABLE 6 for all ports and is compared relative to different standards. We noted that all the power densities satisfy the safety guidelines. The SAR values and power density values are recalculated for the smartphone with housing.
For more investigation of the proposed MIMO antenna with the human head and hand, the antenna performance in terms of the S-parameters and the radiation patterns are introduced as shown in Figure 29 and Figure 30, respectively. A server computer with Xeon-Gold 16 cores processor and 128 GB RAM is used to simulate the smartphone with the human head and hand over the required band. One can be observed that the human head and hand have more effect on the matching of the VP antennas (ports 1, 4, 5, and 8) than that of the HP antennas. This variation occurs because the VP antennas are aligned to the length of the mobile phone that has a large area that is touched with the human than that of the antennas that are aligned to the width of the mobile (HP antennas). Even with the changes in the matching of VP antennas but they still have a good impedance bandwidth that covers more than 4 GHz with reference of -10 dB. The isolation coefficients between MIMO elements are larger than 32 dB. In term of radiation patterns, the MIMO antenna with the human model still achieve the diversity between its ports and achieves high gain.
C. COMPARISON TABLE 7 lists two comparison sections; the first section makes a comparison between the proposed antenna and the referenced dual-polarized antennas and the second section VOLUME 8, 2020 makes a comparison between the proposed antenna and the referenced MIMO antennas of smartphones One can notice that the dual-polarization in [52]- [56] is achieved based on multi-layers and complex feeding structures. However, the antenna in [57] is designed on a single layer but it has a very complicated feeding network to achieve the dualpolarization. Moreover, the antennas in [53], [57] are with low gain. The proposed dual-slot antenna features the benefits of high gain and wide bandwidth, high isolation, high cross-polarization, low profile and compatibility with dualpolarization for 5G applications compared to previous works. On the other hand, the eight-element MIMO antenna for smartphones are introduced in [21], [31], [58], [59] but all the antennas that are introduced in these papers have thick height except the antenna in [31] that has a thickness of 0.8 mm.
The study of the antenna effect on the human body is not introduced in the aforementioned MIMO antennas. Furthermore, these papers achieve low efficiency and low isolation between its ports except the antenna that is introduced in [59] which achieves high efficiency (90%) and accepted isolation between its ports (20 dB). Otherwise, the proposed antenna achieves high isolation between its ports, high gain, high efficiency, and very thin thickness compared to all aforementioned antennas. The proposed work introduces a comprehensive study for all environment of the smartphones and its effect on the antenna performance and vice versa.

VII. CONCLUSION
In this paper, a dual-polarized MIMO antenna with eight elements is introduced for 5G smartphone. The MIMO configuration is based on the diversity between elements. The dual-polarization antenna is introduced to overcome the high attenuation in the 5G communication systems and to give high data rates. Furthermore, the orthogonal-polarization between the antenna ports is used to achieve high isolations between antenna ports. The antenna achieves a good matching bandwidth more than 3 GHz at center frequency of 28GHz. The antenna is combined with MTS to increase its gain and bandwidth. The MTS is analyzed by CMT and all the parameters are investigated. The antenna is fabricated and measured. The electromagnetic exposures into the human model from the proposed antenna at 28 GHz are investigated and analyzed in terms of SAR and power density. High isolation, low profile, low complexity, compact size, high efficiency, high gain, high cross-polarization are achieved in the proposed antenna.   Egypt, in 1968Egypt, in , 1972Egypt, in , and 1975, respectively. She was nominated as an Assistant Professor, an Associate Professor, and a Professor, in 1975Professor, in , 1980Professor, in , and 1985, respectively. In 1989, she was appointed as the President of the Electronics Research Institute (ERI), Cairo, Egypt, a position she held for about ten years. She was the Head of the Microstrip Department, ERI, from 1999 to 2006. She is currently with the Microstrip Department, ERI. She has focused her research on microwave circuit designs, planar antenna systems, and nonreciprocal ferrite devices, and recently on EBG structures, UWB components and antenna and RFID systems. She acts as a single author and as a coauthor on more than 260 research articles in highly cited international journals and in proceedings of international conferences in her field. She supervised more than 80 M.Sc. and Ph.D. thesis. She was PI of more than 30 projects funded from different funding agencies. She has participated in many research projects at the national and international levels under the Egypt-NSF-USA joint funds program, the European Committee Programs FP7 program, NTRA, STDF, ITEDA, and so on. She is also a reviewer for many international societies. She was the authors of published books and chapter in Books in her field of interest. She also participated in many patents.
HADIA S. EL-HENNAWY (Member, IEEE) received the B.Sc. and M.Sc. degrees from Ain Shams University, Cairo, Egypt, in 1972 and 1976, respectively, and the Ph.D. degree from Technische Universitat Braunschweig, Germany, in 1982. In 1982, she returned to Egypt and joined the Electronics and Communications Engineering Department, Ain Shams University, as an Assistant Professor. She was nominated as an Associate Professor, in 1987, and then a Professor, in 1992. In 2004, she was appointed as the Vice-Dean for graduate study and research. In 2005, she was appointed as the Dean of the Faculty of Engineering, Ain Shams University. She has been the Head of the Microwave Research Laboratory, Ain Shams University, since 1982. She has published more than 100 journal and conference papers, and supervised more than 50 Ph.D. and M.Sc. students. She has focused her research on microwave circuit design, antennas, microwave communication, and recently wireless communication. She is a member of the Industrial Communication Committee in the National Telecommunication Regulatory Authority (NTRA), Educational Engineering Committee in the Ministry of Higher Education, and Space Technology Committee in the Academy of Scientific Research. She was the Editor in-Chief of the Faculty of Engineering, Ain Shams University, Scientific Bulletin, from 2004 to 2005. She is deeply involved in the Egyptian branch activities.